RF Module Design - [Chapter 4] Transceiver Architecture

RF Transceiver Module Design
Chapter 4
RF Transceiver Architectures
李健榮助理教授
Department of Electronic Engineering
National Taipei University of Technology

Outline
• General Considerations
• Frequency Conversion
• Receiver Architectures
Heterodyne Receiver
Direct-Conversion Receiver (DCR)
Image-Reject and Low-IF Receiver
• Transmitter Architectures
Direct-Conversion Transmitter (DCT)
Heterodyne and Sliding-IF Transmitter
Open-loop and Closed-loop PLL-based Transmitter
Envelope Tracking and Envelope Following Transmitter
Polar Transmitter
Department of Electronic Engineering, NTUT2/110

Front-end General Considerations
• TX: Adjacent channel leakage
• RX: Rejection of inband and out-of-band interference
BPF
Power Amplifier (PA)
Transmitted Channel
Adjacent Channels
ω
BPF
Low Noise
Amplifier (LNA) Bandpass Filter
Response
Adjacent
Channel
Alternate Adjacent
Channel
1f
f

Interferer Suppression
• High linearity to accommodate interferes without experiencing
compression or significant intermodulation. Filtering the
interferer can relax RX linearity requirements.
• BPF high selectivity is required for near channel rejection.
• Variable BPF is required for different carrier frequencies, and
it is difficult.
900 900.4
( )MHzf
20 dB
35 dB
BPF
Response
Hypothetical filter to suppress an interference

Channel-Selection Filter
• All of the stages in the RX chain that precede channel-
selection filtering must be sufficiently linear to avoid
compression or excessive intermodulation
• Since channel-selection filtering is extremely difficult at the
input carrier frequency, it must be deferred to some other point
along the chain where the center frequency of the desired
channel is substantially lower and hence the required filter Q’s
are more reasonable.

Band-Select Filter
• A band-select filter selects entire RX band and reject out-of-
band interferers, thereby suppressing components that may be
generated by users that do not belong to the standard of
interest.
• Trade-off between selectivity and in-band loss (higher-order
filtering sections and arise NF).
BPF
LNADesired
Channel
Receive Band
f
f
Band-selection filtering

TX-RX Feedthrough
• TX leakage in a CDMA transceiver (full duplex). The RX
must meet difficult linearity requirements.
• A BPF following the LNA can alleviate the leakage.
Duplexer
−20 dBm
LNA
PA
1 W (+30 dBm)
−50 dB
Duplexer
LNA
PA
−50 dB
f
f
TX Leakage
f
BPF Response
BPF
10dB/div.
20 MHz/div.
1f2f
TX Band RX Band
50 dB
30 dB

Frequency Conversion (I)
• Recall Chapter 1 (double sideband amplitude modulation)
( ) ( )cos2m cs t A t f tπ=t( ) ( )BBs t A t=
f
f
cf0 Hzcf−
0 Hz
USBLSB
USBLSBLSBUSB
cos2 cf tπ
“real signal”
Real signal
f
0 Hz
Complex conjugate
USBLSB
1f1f−
cos2 cf tπ
0 Hz cfcf−
USBLSBLSBUSB
IF cf f+c IFf f−IF cf f−c IFf f− −
Double sideband (DSB)

Frequency Conversion (II)
• Recall Chapter 1 (linear modulation)
• Yes, a modulated signal sm(t) is a real signal.
( ) ( ) ( ) ( )1 12 2
2 2
j t j tj f t j f tA t A t
e e e e
φ φπ π− −
= +
( ) ( ) ( )( )1cos 2ms t A t f t tπ φ= +
( ) ( )
{ }12
Re
j t j f t
A t e e
φ π
= ⋅
f
1f0 Hz1f−
“complex”“complex” “real”
Complex conjugate
( )I t
1cos tω
1sin tω−
( )Q t
( )ms t
Real signal
Complex envelope

Frequency Conversion (III)
0 Hz 2f2f−
0 Hz 2f2f−
USBLSBLSBUSB
Real signal
f
0 Hz
Complex conjugate
USBLSB
1f1f− 1 2f f+2 1f f−1 2f f−2 1f f− −
2cos2 f tπ
RFIF
( )I t
cos IFtω
sin IFtω−
( )Q t
( )IFs t
Modulated signal (real signal)
f
0 Hz
USBLSB
IFfIFf−
cos2 cf tπ
RF 0 Hz cfcf−
USBLSBLSBUSB
IF cf f+c IFf f−IF cf f−c IFf f− −
mixing
upconversion
upconversion
IF
LO
LO By filtering, you can choose only USB or LSB
transmission, which is call the single-sideband
(SSB) transmission.

Frequency Conversion (IV)
0 Hz 2f2f−
0 Hz 2f2f−
Real signal
f
0 Hz
Complex conjugate
1f1f− 1 2f f+1 2f f−2 1f f−2 1f f− −2cos2 f tπ
IFRF
downconversion
2 1f f<
2f2f−
0 Hz 2f2f−
0 Hz 2f2f−
Real signal
f
0 Hz 1f1f− 1 2f f+2 1f f−1 2f f−2 1f f− −2cos2 f tπ
IFRF
downconversion
2 1f f>
2f2f−
High-side injection
Low-side injection
LO
LO

Receiver Architecture
• Basic Heterodyne Receiver
• Modern Heterodyne Receiver
Hetero-dyne
Different-freq. Mixing

Basic Heterodyne Receivers (I)
• Translating the desired channel to a much lower center
frequency to permit a channel-selection filtering with a
reasonable Q.
ω
inωinω− 0
ω
LOωLOω− 0
Downconversion by mixing
RF input
inω
ω
Mixer
0 cos LOA tω
vLPF IF Output
in LOω ω− − in LOω ω− + 0 in LOω ω+in LOω ω−
Filtered-outFiltered-out
LO
( ) ( )
1 1
cos cos cos cos
2 2
in LO in LO in LOt t tω ω ω ω ω ω⋅ = + + −
Low freq.High freq.
Two IF frequencies:

Basic Heterodyne Receivers (II)
• Use of LNA to reduce noise
• Variable IF:
• Constant IF:
Mixer
0 cos LOA tω
vLPF IF Output
RF input
LNA
IFj RFj LOf f f= − (Constant LO freq. and variable IF freq.)
IF RFj LOjf f f= − (Variable LO freq. and constant IF freq.)
Precise LO freq. and steps provided by a “frequency synthesizer”
Constant IF approach is more common to simplify the design of IF path; e.g., it does not
require a variable-frequency channel selection filter.
LO

Basic Heterodyne Receivers (III)
• Constant-LO downconversion mixing
• Constant-IF downconversion mixing
1RFf
f f
f
1LOf
IFf0
1RFf
f f
f
LOf
1IFf0 2RFf
f f
f
2IFf0
LOf
2RFf
f f
f
IFf0
2LOf

• While each wireless standard impose constrains upon the
emissions by its own users, it may have no control over the
signals in other bands. The image power can therefore be
much higher than that of the desired signal, requiring proper
“image rejection.”
Image Problem in Heterodyne RX
cos LOtω
vLPF
Desired signal
Image
inω imω
ω
IFω
ω
IFω IFω
LOω
ω
High-side injection
( ) ( )cosd d dA t t tω φ +   ( ) ( )cosim im imA t t tω φ +  
( ) ( ) ( ) ( ) ( ) ( ) ( )
1 1
cos cos
2 2
IF d LO d LO d d LO d LO dx t A t A t t A t A t tω ω φ ω ω φ   = + + − − +   
( ) ( ) ( ) ( ) ( ) ( )
1 1
cos cos
2 2
im LO im LO im im LO im LO imA t A t t A t A t tω ω φ ω ω φ   + + + − − +   

Downconverted Spectrum (I)
1ω−2ω− 1ω+ 2ω+0
ω
ω
ω
0 LOω+LOω−
0
1ω−2ω− 1ω+ 2ω+0
ω
ω
ω
0 LOω+LOω−
0
Downconversion for 1LOω ω< Downconversion for 2 1LOω ω ω> >

Downconverted Spectrum (II)
1ω−2ω− 1ω+ 2ω+0
ω
ω
ω
0 LOω+LOω−
0
1ω−2ω− 1ω+ 2ω+0
ω
ω
ω
0 LOω+LOω−
0
Downconversion for 2 1LOω ω ω> > Downconversion for 2LOω ω>
1 2
2
LO
ω ω
ω
+
=

• The most common “image rejection “ approach is to precede
the mixer with an “image-rejection filter.”
• The filter exhibits a relatively small loss in the desired band
and a large attenuation in the image band, two requirements
that can be simultaneously met if 2ωIF is sufficiently large.
• A filter with high image rejection typically appears between
the LNA and the mixer to lower the noise contribution to the
RX NF (The NF increases while the filter precedes the LNA).
Image Rejection
Image Reject Filter
inω imω
ω
2 IFω
cos LOtω
v
Image
Reject
Filter
LNA

Image Rejection v.s. Channel Selection
Image Reject Filter
inω imω
ω
2 IFω
cos LOtω
v
Image
Reject
Filter
LNA
v
Channel
Select
Filter
Desired channel
Interference
Image
Channel Select Filter
(high-Q needed)
IFω
ω
0
0
ω
IFω
inω imω
2 IFω
ω
High IF
Low IF
• If the IF is high, the image can be
suppressed but complete channel
selection is difficult, and vice versa.

Image Noise Increases Noise Figure
• Even in the absence of interferes, the thermal noise produced
by the antenna and the LNA in the image band arrives at the
input of the mixer. The thermal noises in the desired channel
and image band are downconverted to IF (unless the LNA has
a limited bandwidth to suppresses the noise in the image band).
LOω
LNA
inω
Thermal Noise
LOω inω
ω
2 in LOω ω−
ω

Dual IF Receiver (I)
• The concept of heterodyning is extended to multiple
downconversions, each followed by filtering and amplification,
to resolve the trade-off between “image rejection” and
“channel selection.”
• This technique performs partial channel at progressively lower
center frequencies, thereby relaxing the Q required of each
filter.
1LOω
vBPF2
LNA
vBPF1 vBPF3
2LOω
vBPF4
Band
Select
Filter
Image
Reject
Filter
RF Mixer
MX1
Channel
Select
Filter
IF Mixer
MX2
Channel
Select
Filter IF Amp.
A C E GB D F H

Dual-IF Receiver (II)
1LOω
vBPF2
LNA
vBPF1 vBPF3
2LOω
vBPF4
Band
Select
Filter
Image
Reject
Filter
RF Mixer
MX1
Channel
Select
Filter
IF Mixer
MX2
Channel
Select
Filter IF Amp.
A C E GB D F H
Department of Electronic Engineering, NTUT
Desired Channel Image
fA
C
E
G
B
D
F
H
BPF1
BPF2
Image
Image
BPF3
BPF4
f
f
f f
f
f
f
23/110

Mixing Spurs (I)
• In practice, the mixing is the multiplication of the RF input by
all harmonics of the LO. Thus the RF mixer produces
components at and IF mixer, where
m and n are integers.
• For the desired signal, only is of interest. But if
an interferer, , is downconverted to the same IF, it corrupts
the signal; this occurs if .
1in LOmω ω± 1 2in LO LOm nω ω ω± ±
1 2in LO LOω ω ω− −
intω
int 1 2 1 2LO LO in LO LOm nω ω ω ω ω ω± ± = − −

Mixing Spurs (II)
• An example of a 2.4 GHz dual conversion RX :
1 1.98 GHzLOω =
LNA
vBPF
2 400 MHzLOω =
420 MHz2.4 GHz
20 MHz
2.76GHz
2.4 GHz
2.8GHz
4.38 GHz
f
Received Spectrum
820 MHz
780 MHz
2800 MHz
2760 MHz
22 800 MHzLOω =
12 3.96 GHzLOω =
4380 MHz
20 MHz
420 MHz
2 400 MHzLOω =
20 MHz

Modern Heterodyne Receivers
ω
ω
0
0
1IFω− 1IFω+
2LOω+2LOω−
ω
ω
Desired Channel
Interferer
0
0
• Zero Second IF: Avoid secondary image (assume no interferers are
downconverted as an image to a zero center frequency).
• Interferer appears in the adjacent channel
2LOω
ω
ω
1IFω ω
02 1LO IFω ω=
2 1LO IFω ω=

Signal Becomes its Own Image
• For symmetrically-modulated signal:
• For asymmetrically-modulated signal:
LOf
f
f
0 f
LOf
( )S f ( )LOS f f−
vVCO
( )BBx t
t f
cf
1IFω+1IFω− 0
ω
ω
0
same information on both side of the carrier
downconversion
Corruption occurs if the signal spectrum is asymmetric

Avoid Self-corruption of Asymmetric Signals (I)
• One can downconvert signal to an IF equal to half of the signal
bandwidth to avoid self-corruption of a signal with asymmetric
spectrum.
1IFω+1IFω− 0
ω
ω
0
BWω
2
BWω
+
2
BWω
−
1
2
BW
IF
ω
ω ≥

Avoid Self-corruption of Asymmetric Signals (II)
• Zero second IF with quadrature downconversion.
( )IFx t
( ),BB Ix t
( ),BB Qx t
2cos LO tω
2sin LO tω
2 1LO IFω ω=
Quadrature
baseband signal
Though xBB,I(t) and xBB,Q(t) exhibit identical spectra, they are separated in phase
and together can reconstruct the original information

Zero Second IF Heterodyne RX
• Zero second IF heterodyne RX with quadrature downconverison
( )IFx t
( ),BB Ix t
( ),BB Qx t
2cos LO tω
2sin LO tω
1LOω
vBPF
LNA
No image rejection filter
LNA/mixer interface can be optimized (need not 50 Ohms) for gain, noise, and
linearity with little concern for the interface impedance values.
The lack of image rejection filter requires careful attention to the interferers in
the image band, and dictates a narrow-band LNA design (suppress image
noise).
No channel-selection filter is shown at the first IF, but some “mild” on-chip
BPF is usually inserted to suppress out-of-band interferers.

Sliding-IF Heterodyne RX (I)
( )IFx t
( ),BB Ix t
( ),BB Qx t
2,ILO
1LOω
vBPF
LNA
vLO1 v2÷
2,QLO
t
2,ILO
2,QLO
1LO
90
RF Input
1st LO
1st IF
2nd IF
inf
f
f
f
f
1LOf
1in LOf f−
1
1
2
LO
in LO
f
f f− −
For an input band [f1, f2], the LO
must cover a range of [(2/3)f1, (2/3)f2].
1
1
2
LO
LO in
f
f f+ = 1
2
3
LO inf f=
The 1st IF is not constant, because
1
1
3
IF in LO inf f f f= − =

Sliding-IF Heterodyne RX (II)
• As fin varies from f1 to f1, fIF1 goes from f1 /3 to f2 /3 (slide IF).
RF Range
LO Range
1st IF Range
f
1f 2f
f
f
1
2
3
f 2
2
3
f
1
1
3
f 2
1
3
f
( )
2 1
2 1
1 1
3 3% *100%
1 1 1
2 3 3
IF
f f
BW
f f
−
=
 
+ 
 
( )
( )
2 1
1 2
% *100%
1
2
RF
f f
BW
f f
−
=
+

Sliding-IF Heterodyne RX (III)
• Image band of the sliding-IF heterodyne RX
Image Band RF Band
LO Band
1
1
3
f 2
1
3
f 1f 2f
f
f
1
2
3
f 2
2
3
f

Direct-Conversion Receivers
• Direct-conversion receiver (DCR) is also called the “zero-IF”,
or the “homodyne” receiver.
• As mentioned previously, downconversion of an asymmetric-
modulated signal to a zero IF leads to self-corruption unless
the baseband signals are separated by their phases.
I
Q
cos LOtω
sin LOtωvBPF
LNA
vLPF
vLPF
inω
LO inω ω=

DCR Advantages
• The absence of an image greatly simplifies the design process.
• Channel selection is performed by low-pass filters, which can
be realized on-chip as active circuit topologies with relatively
sharp cut-off characteristics.
• Mixing spurs are considerably reduced in number and hence
simpler to handle.
• The LNA/mixer interface can be optimized for gain, noise, and
linearity without requiring a 50- impedance.

DCR Issues − LO Leakage
• A DCR emits a fraction of its LO power from its antenna, and
the LO emission is undesirable because it may desensitize
other receivers operating in the same band. Typical acceptable
values range from −50 to −70 dBm (measured at the antenna).
• In heterodyne receivers, since the LO frequency falls outside
the band, it is suppressed by the front-end band-select filters in
both the emitting receiver and the victim receiver.
Pad
LNA
LO
Substrate
LO
LNA
LO leakage Cancellation of LO leakage by symmetry

DCR Issues − DC Offset
• The leakage causes “LO self-mixing” at the mixer to produce a
dc component in the baseband (because multiplying a sinusoid
by itself results in a dc term). The zero second IF architecture
also suffers from this issue.
• LO leakage yields a very large dc offset due to the high gain of
the receiving chain, and this saturates the baseband circuits
(degrades the dynamic range), prohibiting signal detection.
• Time-varying dc offset
RF LOV kV+
Pad
LNA
IF DCV V+

Effect of DC Offset in Baseband Chain
• Since and , thus . Amplified
by another 40 dB, this offset reaches 1-V at the baseband
output.
1 31.6vA = ( )632 2 VleakV µ= 10 mVdcV =
cos LOtω
LPF
sin LOtω
LNA
1 30 dBvA = 2 40 dBvA =
0 cos inV tω
( )cosbb in LOV tω ω−
1 0bb vV A V=
cosleak LOV tω
1dc v leakV A V=

Leakage of Quadrature Phases of LO
• The dc offset measured in the baseband I and Q outputs are
often unequal.
LO
LNA
( )cosleak LO leakV tω φ+
( ), cosdc I leak LO leak circuitV V Vα φ φ= +
( ), sindc Q leak LO leak circuitV V Vα φ φ= − +

Cancelling DC Offset
LNA
cos LOtω
LPF
1C
1R
bv
1A
Baseband SignalHPF Response
1f− 1f+0
f
• Using a HPF (ac coupling) removes dc offset but also removes
a fraction of the signal’s spectrum near zero frequency, thereby
introducing distortion.
• As a rule of thumb, the corner frequency of the HPF must be
less than 1/1000 of the symbol rate for negligible distortion.
This may require very large capacitance and thus difficult to
integrate on chip (especially for low symbol rate). For the slow
response to transient inputs (LO switch, LNA gain change), ac
coupling is rarely used in today’s receivers.

DCR Issues − Even-order Distortion
• DCRs are additionally sensitive to even-order nonlinearity in
the RF path (IM2 falls around DC to corrupt the desired signal
and mixer feedthrough), and so are the heterodyne
architectures having a second zero IF .
Beat
Component
cos LOtω
1 2ω ω−
Feedthrough
LNA
Desired
Channel
Interferers
0
ω
0
ω
1ω
ω
2ω

Mixer Feedthrough – Simple Mixer
inV outV
LO
1R
( ) ( ) ( ) ( ) ( ) ( )
1 1
2 2
out in in inV t V t S t V t S t V t
 
= ⋅ = − + ⋅  
where is the RF input and is the Ideal LO toggling between 0 and 1 with 50%
duty cycle, and
( )
1
2
S t
 
−  
DC-free square wave
( )
1
2
inV t ⋅ is the RF feedthrough to the output
( )inV t ( )S t
represents a

Mixer Feedthrough – Differential Mixer
• If the output is sensed differentially, the RF feedthrough Vout1(t)
and Vout2(t) are cancelled while the signal components add.
• This cancellation is sensitive to asymmetrics, e.g., if the
switches exhibits a mismatch between their on-resistance, then
a net RF feedthrough arises in the differential output.
( ) ( ) ( )1out inV t V t S t= ⋅
LO
LO
1R
1R
1outV
2outV
inV
( )S t
( )1 S t−
( ) ( ) ( )2 1out inV t V t S t= ⋅ −  

Even-order Distortion (I)
• The 2nd order nonlinear effect cause the beat amplitude which
grows with the square of the amplitude of the input tones.
(log scale)
(log scale)
inA
IIP2A
IIP2A
1Aα
2
2 Aα
( ) ( ) ( )2
1 2out in inV t V t V tα α= +
( ) ( ) ( )2 2
1 1 2 2 1 2 2 1 2cos cos cos cosA t t A t A tα ω ω α ω ω α ω ω= + + + + + +⋯
( ) 1 2cos cosinV t A t A tω ω= +
• Since the net feedthrough of the
beat depends on the mixer and
LO asymmetries, the beat
amplitude measured in the
baseband depends on the derive
dimensions and the layout and is
therefore difficult to formulate.
44/110

Even-order Distortion (II)
• Even-order distortion may manifest itself even in the absence
of interferers. Suppose in addition to frequency and phase
modulation, the received signal also exhibits amplitude
modulation.
• Both of the terms and are low-pass signals and,
like the neat component, pass through the mixer with finite
attenuation, corrupting the downconverted signal.
( ) ( ) ( )0 cosin cx t A a t t tω φ=  +   +    
( ) ( ) ( )2
1 2out in inV t V t V tα α= +
( ) ( ) ( )
( )2 2 2
2 2 0 0
1 cos 2 2
2
2
c
in
t t
x t A A a t a t
ω φ
α α
+  +   = + + 
( )2 0A a tα ( )2
2 2a tα

DCR Issues − Flicker Noise (I)
• Linearity requirements limit the cascaded LNA/mixer gain, the
downconverted signal in a DCR is still relatively small and
hence susceptible to noise in the BB circuits. Since the signal
is centered around zero frequency, it can be substantially
corrupted by the flicker noise.
• The mixers themselves may also generate flicker noise at their
output.
(logscale)
f
BWfCf
1000
BWf
( )1 fS f
thS
,
1
2
BW RF BWf f=
1 fS
f
α
= th
c
S
f
α
= th cS fα = ⋅
Assume noise components below fBW/1000 are unimportant
( )1 0.001
5.9 ln
BW
BW
f
c
n BW c th c th BW thf
BW
f
P df f f S f S f S
f f
α  
= + − = + + 
 
∫
If no flicker noise 2n BW thP f S≈
Flicker noise penalty 1
2
n
n
P
P

DCR Issues − Flicker Noise (II)
• In a good design, the thermal noise at the end of the baseband
chain arises mostly from the noise of the antenna, the LNA,
and the mixer. Thus, a higher front-end gain directly raises Sth,
thereby lowering fc and hence the flicker noise penalty.
• Flicker noise penalty:
An 802.11g RX with fc of 200 kHz:
10 MHzBWf =
1
2
1.04n
n
P
P
=
A GSM RX with fc of 200 kHz:
1
2
16.4n
n
P
P
=
(logscale)
BWf200100
( )1 fS f
( )kHzf
thS
Downconverted
GSM Channel
Effect of flicker noise on a GSM channel
Flicker noise makes it difficult to employ DCR for a narrow channel bandwidth.
In such cases, the “low-IF” architecture proves a more viable choice.

DCR Issues − I/Q Mismatch
• DCR require 90o shift of the RF signal and this generally
entails severe noise-power-gain trade-offs.
I
Q
LOV
vLPF
vLPF
90
RFV
I
Q
LOV
vLPF
vLPF
RFV
90
Shift of RF signal or LO waveform by 90o

I/Q Mismatch (I)
• Errors in the 90o phase shift circuit and mismatches between
the quadrature mixers result in imbalance in the amplitudes
and phases of the baseband I and Q outputs.
• The BB stages themselves may also contribute mismatches.
I
Q
LOV
vLPF
vLPF
RFV
90
Phase and Gain Error
Phase and
Gain Error
Phase and
Gain Error
Phase and Gain Error

I/Q Mismatch (II)
• I/Q mismatches tend to be larger in DCRs than in heterodyne
topologies, because:
Propagation of a higher frequency experiences greater mismatches
LO quadrature phases suffer from greater mismatches at higher frequencies
cos LOtω
sin LOtω
LNA
5 GHz
10 ps 18 @5 GHzT∆ = ⇒
( )IFx t
( ),BB Ix t
( ),BB Qx t
LNA
vLO1 4÷
5 GHz
4 GHz
1 GHz
10 ps 3.6 @1 GHzT∆ = ⇒
DCR Heterodyne RX

I/Q Mismatch – QPSK Example (I)
( ),BB Ix tvLPF
vLPF
( )inx t 90 ( )LOx t
2
θ
+
2
θ
−
1
2
ε
+
1
2
ε
−
( ),BB Qx t
( ) cos sinin c cx t a t b tω ω= +
, 1a b = ±
( ), 2 1 cos
2 2
LO I cx t t
ε θ
ω
   
= + +   
   
( ), 2 1 cos
2 2
LO Q cx t t
ε θ
ω
   
= − −   
   
( ), 1 cos 1 sin
2 2 2 2
BB Ix t a b
ε θ ε θ   
= + − +   
   
( ), 1 sin 1 cos
2 2 2 2
BB Qx t a b
ε θ ε θ   
= − − + −   
   
The mismatch causes crosstalk between I and Q BB signals.

I/Q Mismatch – QPSK Example (II)
I
Q
t
t
I
Q
Ideal
I
Q
t
t
I
Q
Ideal
Only amplitude mismatch : 0, 0ε θ≠ =
Only phase mismatch : 0, 0ε θ= ≠
( ), 1
2
BB Ix t a
ε 
= + 
 
( ), 1 cos
2 2
BB Qx t b
ε θ 
= − 
 
( ), cos sin
2 2
BB Ix t a b
θ θ
= −
( ), sin cos
2 2
BB Qx t a b
θ θ
= − +

Correction of I/Q Mismatch
cos LOtω
sin LOtω
LPF
LPF
LNA
I
Q
I
Q
Phase
Mismatch
Amplitude
Mismatch
t
LPF
LPF
ADC
ADC
Logic
φ
φ
cos LOtω
sin LOtω
LNA
• Calibration of quadrature phase and gain either at power up or
continuously is usually needed for high performance system.
Test signal
Analog adaption
Digital adaption is more popular
in nowadays systems

Mixing Spurs
• Unlike heterodyne systems, DCRs rarely encounter corruption
by mixing spurs. This is because, for an input frequency f1 to
fall in the baseband after experiencing mixing with nfLO, we
must have f1 ≈ nfLO.
• Since fLO is equal to the desired channel frequency, f1 lies far
from the band of interest and is greatly suppressed by the
selectivity of the antenna, the band-select filter, and the LNA.
• The issue of LO harmonics does manifest itself if the receiver
is designed for a wide frequency band (greater then two
octaves). Examples include TV tuners, “software-defined
radios,” and “cognitive radios.”

Image-Reject Receivers
• “Image-reject” architectures are another class of receivers that
suppress the image without filtering, thereby avoiding the
trade-off between image rejection and channel selection.
• Benefits from a 90o shifter (Hilbert transform, −j for f > 0, +j for f <0)
Re
Im
Re
Im
Re
Im
2
A
j+
2
A
2
A
2
A
j−
cω−
cω+
ω
( )cos
2
c cj t j t
c
A
A t e eω ω
ω −
= +
Illustration of 90o phase shift for a cosine
( ) ( )9090
cos 90
2
cc
j tj t
c
A
A t e e
ωω
ω
− −− − = + 
 
2 2
c cj t j tA A
je jeω ω−
= − +
sin cA tω=

Hilbert Transform (I)
• Hilbert transformation pair:
• The transform means a 90 degree phase shift in time domain,
the impulse response of the Hilbert transformation .
( )
( )
( )
ˆ
x
x t d
t
τ
τ
π τ
∞
−∞
=
−∫ ( )
( )
( )
ˆx t
x d
t
τ τ
π τ
∞
−∞
= −
−∫
( ) 1h t tπ=
( )
1
h t
tπ
=( )x t ( )ˆx t
90( )x t ( )ˆx t

Hilbert Transform (II)
• Simple relation between sine and cosine functions:
• It simply shows that if we want to make a transform between
cosine and sine waveforms, a 90 degrees phase shift is
required.
• From Euler’s formula:
( ) ( )cos 90 sint tω θ ω θ+ − = + ( ) ( )sin 90 cost tω θ ω θ+ − = − +and
0 0
0cos
2
j t j t
e e
t
ω ω
ω
−
+
=
0 0
0sin
2
j t j t
e e
t
j
ω ω
ω
−
−
=
( ) ( )0 0
1
2
δ ω ω δ ω ω− + +  
( ) ( )0 0
1
2 j
δ ω ω δ ω ω− − +  
F.T.
F.T.

Hilbert Transform (III)
• We like to find a transfer function, which is able transfer the
cosine to since and since to cosine function.
( ) ( ) ( ) ( ) ( )0 0 0 0
1 1
2 2
H j jδ ω ω δ ω ω ω δ ω ω δ ω ω− + + ⋅ = − − − +      
( ) ( ) ( ) ( ) ( )0 0 0 0
1 1
2 2
H j
j
δ ω ω δ ω ω ω δ ω ω δ ω ω   − − + ⋅ = − − + +   
( ) ( )sgnH j jω ω= − ⋅
( )
1 , 0
sgn 0 , 0
1, 0
ω
ω ω
ω
>

= =
− <
( ) ( )
0
0
1 1
sgn
2 2 2
j t j t j tj j
h t j e d e d e d
t
ω ω ω
ω ω ω ω
π π π π
∞ ∞
−∞ −∞
= − ⋅ ⋅ = − =∫ ∫ ∫
cosine
sine negative cosine
0 0ω ω= > ( ) ( ) ( )0
1 1
0 0
2 2
H j jδ ω δ⋅ = −
0 0ω ω= − < ( ) ( ) ( )0
1 1
0 0
2 2
H j jδ ω δ⋅ − =
0 0ω ω= = ( ) ( ) ( ) ( ) ( )
1 1
0 0 0 0 0
2 2
H jδ δ δ δ+ ⋅ = − −       ( )0 0H =
( )H j jω = −
( )H j jω =
:
:
:
sine
phase
f
90+
90−

90o Phase Shift (I)
• 90o phase shift for a modulated signal
Re
Im
( )X ω
ω
cω+
cω−
( )X ω( )jX ω
( )jX ω−
( ) ( ) ( )cos cx t A t t tω φ=  +  
( ) ( )
( ) ( )( ) ( )( ) ( ) ( )( ) ( )( )90 90
cos 90
2 2
c c c c
j t t j t t j t t j t t
c
A t A t
A t t t e e je je
ω φ ω φ ω φ ω φ
ω φ
+ − − + − + − +    + − = + = − +      
( ) ( )sin cA t t tω φ=  +  
( ) ( ) ( )90
sgnX X jω ω ω= −  

90o Phase Shift (II)
Re
Im
2
A
j+
2
A
2
A
j−
cω−
cω+
ω
sin
2
c cj t j t
c
e e
A t A
j
ω ω
ω
−
−
=
2
A
−
sin cjA tω
Re
Im
2
A
2
A
cω−
cω+
ω
cos ctω
Re
Im
A
cω−
cω+
ω
cos sinc ct jA tω ω+
• Plot the spectrum of cos sinc cA t jA tω ω+
(SSB spectra)
sine is the Hilbert transform of the cosine
To get a SSB spectra: (1) real signal (2) its Hilbert transform (3) = (1)+j(2)
60/110

90o Phase Shift (III)
Re
Im
( )S ω
ω
cω+
cω−
( )S ω( )jS ω
( )jS ω−
Re
Im
( )S ω
ω
cω+
cω−
( )S ω−
( )jS ω
( )jS ω−
Re
Im
( ) ( )ˆS jSω ω+
ω
cω+
• A narrowband signal with a real spectrum is shifted by 90o
to produce . Plot the spectrum of which is called
the “analytic signal,” or the “pre-envelope” of .
( )s t
( )ˆs t ( ) ( )ˆs t js t+
( )s t
( )ˆS ω ( )ˆjS ω

90o Phase Shift (IV)
• Use RC-CR network to implement the 90o phase shifter
1C
1R
1R
1C
1outV
2outV
inV
HPFH
LPFH
1
1
2
1 1
1
R C
ω
1
1 1tan
2
HPFH R C
π
ω−
∠ = −
1
1 1tanLPFH R C ω−
∠ = −
ω
2
π
2
π
−
0
2
π
( ) 1 1 1
1 1 1
out
HPF
in
V R C s
H s
V R C s
= =
+
( ) 2
1 1
1
1
out
LPF
in
V
H s
V R C s
= =
+
We can consider Vout2 as
the Hilbert transform of
Vout1 at frequencies close
to (R1C1)−1
Amplitude response
Phase response
62/110

Quadrature Downconversion (I)
• Quadrature downconversion translate the spectrum to a
nonzero IF as a 90o phase shifter.
RFV
IFI
cos LOtω
sin LOtω
IFQ
0cω− cω+
ω
0LOω− LOω+
ω
1
2
+
1
2
+
1
2
+
1
2
+
0IFω− IFω+
ω
0cω− cω+
ω
2
j
+
2
j
−
LOω+
0LOω−
ω
0IFω−
IFω+
ω
2
j
−
2
j
+
High-side injection

Quadrature Downconversion (II)
• Quadrature downconversion translate the spectrum to a
nonzero IF as a 90o phase shifter.
2
j
+
2
j
−
LOω+
0LOω−
ω
0
IFω−
IFω+
ω
2
j
−
2
j
+
1
2
+
1
2
+
0IFω− IFω+
ω
Low-side injection
RFV
IFI
cos LOtω
sin LOtω
IFQ
0cω− cω+
ω
0cω− cω+
ω
0LOω− LOω+
ω
1
2
+
1
2
+

Distinguish Desired Signal and Its Image
( )sig im IF
I I+
cos LOtω
sin LOtω
( )sig im IF
Q Q+
Re
Im
sigI
IFω−
sigQ
sigQ
sigI
IFω+ ω
Re
Im
imI
imQ
imQ
imI
IFω+ ω
Re
Im
cω+ ω
imω+
imω−
cω−
Signal Components
Image Components
IFω−
Signal: low-side injection
Image: high-side injection
LOω+
LOω−
65/110

Image Reject RX – Hartley Architecture (I)
• Negate image
C
A
LPF
LPF
B
cos LOtω
sin LOtω
90
sigQ
imQ
,90im
Q
,90sig
Q
sigI
imI
IF Output
ω
imω+ cω+cω− imω− 0
sigI
sigQ
imI
imQ
Re
Im
IFω−
IFω+ ω
Re
Im
IFω−
IFω+
ω
Re
Im
IFω−
IFω+ ω
Re
Im
IFω−
IFω+ ω
66/110

Image Reject RX – Hartley Architecture (II)
C
A
LPF
LPF
B
cos LOtω
sin LOtω
90
sigQ
imQ
,90im
Q
,90sig
Q
sigI
imI
IF Output
sigI imI
Re
Im
IFω−
IFω+ ω
Re
Im
IFω−
IFω+ ω
,90sig
Q ,90im
Q
Re
Im
IFω−
IFω+ ω
Re
Im
IFω−
IFω+
ω
Re
Im
IFω−
IFω+ ω
67/110
• Negate image

Image Reject RX – Hartley Architecture (III)
• Realization of 90o phase shift in Hartley receiver
1cos tω
1sin tω
RF Input
LPF
LPF
IF Output
1R
1R
1C
1C

Image Reject RX – Hartley Architecture (IV)
• Downconversion of Hartley receiver output to baseband
LPF
LPF
1cos LO tω
1sin LO tω
90
2sin LO tω
2cos LO tω
I
Q
RF Input

Image Reject RX – Weaver Architecture (I)
A
1cos tω
1sin tω
RF Input
LPF
LPF
2cos tω
2sin tω
LPF
LPF
B
C
D
E
F
−
+
IF Input
ω
imω+ cω+cω− imω− 0
sigI imI
Re
Im
1IFω−
1IFω+ ω
Re
Im
1IFω−
1IFω+ ω
sigQ imQ
Re
Im
1IFω−
1IFω+ ω
Re
Im
1IFω−
1IFω+
ω
70/110
• Negate image

Image Reject RX – Weaver Architecture (II)
A
1cos tω
1sin tω
LPF
LPF
2cos tω
2sin tω
LPF
LPF
B
C
D
E
F
−
+
IF
sigI imI
Re
Im
2IFω−
2IFω+ ω
Re
Im
2IFω−
2IFω+ ω
sigQ imQ
Re
Im
2IFω−
2IFω+ ω
Re
Im
2IFω−
2IFω+ ω
Re
Im
IFω−
IFω+ ω
Low-side inj.
71/110
• Negate image

Image Reject RX – Weaver Architecture (III)
1inω ω−
ω
2ω
2 12 inω ω ω− +
0
First
IF
ω
1 2inω ω ω− −
0
Second
IF
ω
2 12 2inω ω ω− +
1ω inω0
RF
Input
Secondary
Image
Desired
Channel
• Problem of secondary image
72/110

Image Reject RX – Weaver Architecture (IV)
1cos tω
1sin tω
RF Input
LPF
LPF
2÷ ( ),BB Qx t
−
+−
+
( ),BB Ix t
• Double quadrature downconversion Weaver architecture to
produce BB outputs. The second downconverion produces
zero IF to avoid secondary image.
73/110

Low-IF Receiver (I)
cf
GSM Adjacent Channel Spec.
9 dB
ff
0 100 kHz
cf
f
LOf
f
200 kHz
• It is undesired to place image within the signal band because
the overall NF would raise by approximately 3 dB.
• In “low-IF” RXs, the image indeed falls in the band but can be
suppressed by image rejection techniques.
• For a GSM RX, signal would be corrupted by flicker noise in a
zero-IF architecture. The noise penalty can be lower by using
low-IF architecture (attractive for narrow-channel standards).
Moderate IRR is ok.

Low-IF Receiver (II)
2sin tω
LPF
2cos tω
LPF
1R
1R
1C
1C
IF OutputRF Input
Quadrature Phases of
Image and Signal
sin LOtω
LPF
cos LOtω
IF OutputRF Input
LPF
ADC
ADC
90
• Adopt image cancellation technique with low-IF architecture
75/110

Low-IF Receiver (III)
• Low-IF receiver with double quadrature downconverter
sin ctω
cos ctω
IF Output
RF Input
( ),IF Qx t
−
+ +
+
( ),IF Ix t90
76/110

Polar Receiver
• Using the oscillator injection locking technique to accomplish
magnitude and phase extraction of the complex envelope. This
technique was also published to perform envelope elimination
and restoration in the Kahn EER transmitter.
LPF
LPF
ILO1 ILO2
Magnitude
Phase

Transmitter Architectures
• Basic Direct Conversion Transmitter (DCT)
• Modern DCT
• Heterodyne Transmitters
• OOK Transceivers
• Open-loop Phase Modulation Techniques
• Closed-loop Phase Modulation Techniques
• Polar Transmitter

• Quadrature upconverter:
Server as the modulator.
• Power amplifier:
Amplify the signal.
• Matching network :
Provide maximum power delivery to antenna and filter out-of-band components
that result from the PA nonlinearity.
• xBB,I(t) and xBB,Q(t) are generated by BB circuits and hence has a sufficiently
large amplitude, the noise of the mixers is much less critical here than in
receivers.
• A predriver is typically interposed between the upconverter and the PA to serve
as a buffer.
Direct-Conversion Transmitter (DCT)
cos ctω Matching
Network
PA
sin ctω−
( ) ( ) ( )cos cx t A t t tω φ=  +  
( ) ( ) ( ) ( )cos cos sin sinc cA t t t A t t tφ ω φ ω= −
( ) ( ) ( ), cosBB Ix t A t tφ=
( ) ( ) ( ), sinBB Qx t A t tφ=

I/Q Mismatch
• I/Q mismatch in the DCT:
• Constellation:
( ) ( ) ( )1 2cos sinc c c c cx t A A t A tα ω θ α ω= + ∆ + ∆ +
( ) ( )1 2 1cos cos sin sinc c c c c c cA A t A A A tα θ ω α α θ ω = + ∆ ∆ + − + ∆ ∆ 
1 2, 1α α = ±
1 21 cos , 1 1 sinc c
c c
A A
A A
β θ β θ
   ∆ ∆
= + + ∆ = − + ∆   
   
1 21 cos , 1 1 sinc c
c c
A A
A A
β θ β θ
   ∆ ∆
= + + ∆ = − − + ∆   
   
1 21 cos , 1 1 sinc c
c c
A A
A A
β θ β θ
   ∆ ∆
= − + ∆ = + + ∆   
   
1 21 cos , 1 1 sinc c
c c
A A
A A
β θ β θ
   ∆ ∆
= − + ∆ = − + + ∆   
   

I/Q Calibration (I)
• Apply a single sinusoidal to both inputs of the upconverter to
reveal phase mismatch.
It can be shown that the output contains two sidebands of equal amplitudes and
carries an average power equal to
We observe that ε is forced to zero as described above, then .
Thus, the calibration of phase mismatch proceeds to drive this quantity to zero.
cos ctω
sin ctω0 cos inV tω 3outV
+
−
( ) ( ) ( )3 0 01 cos cos cos sinout in c in cV t V t V tε ω ω θ ω ω= + + ∆ −
( )0 cos 1 cos cosin cV tω ε θ ω= + ∆
( ) ( )2 2
3 0 1 1 sinoutV t V ε θ=  + + ∆  
2 2
3 1 sinout outV V θ− = ∆
( )0 cos 1 sin 1 sinin cV t tω ε θ ω− ⋅  + ∆ +  

I/Q Calibration (II)
• Applying a sinusoidal to one BB input while the other is set to
zero for gain mismatch calibration.
The gain mismatch can be adjusted so as to drive this difference to zero.
0 cos inV tω
cos ctω
sin ctω 1outV
cos ctω
sin ctω
0 cos inV tω
2outV
( ) ( ) ( )1 0 1 cos cosout in cV t V t tε ω ω θ= + ⋅ + ∆ ( )
2
2 20
1 0
2
out
V
V t V ε= +
( )2 0 cos sinout in cV t V t tω ω= ⋅ ( )
2
2 0
2
2
out
V
V t =
( ) ( )2 2 2
1 2 0out outV t V t V ε− =

Carrier Leakage (I)
• The analog BB circuitry producing the quadrature signals
exhibits dc offsets, and so does the baseband port of each
upconvertion mixer:
• The upconverter therefore contains a fraction of the
unmodulated carrier, called “carrier leakage”:
( ) ( ) ( )1 2cos cos sin sinout OS c OS cV t A t V t A t V tφ ω φ ω=  +  −  +    
( ) ( ) ( ) 1 2cos cos sinout c OS c OS cV t A t t V t V tω φ ω ω= + + −
( )
2 2
1 2
2
Relative Carrier Leakage OS OSV V
A t
+
=

Carrier Leakage (II)
• RX BB outputs suffer from dc offsets.
• In the presence of carrier leakage, if
the TX power is controlled by
varying BB signals, it is difficult
forThe base station to measure the
actual signal power.
0 2OSV V+ +
Q
I
0 1OSV V+ +0V− 0V+
0V−
0V+
cos ctω
sin ctω
PA
Receiver
Base Station
Baseband
Processor
Carrier Leakage
cω
ω

Reduction of Carrier Leakage
Baseband
Processor
DACQ
DACI
Register ADC
Power
Detector
cos ctω
sin ctω
• The BB swing, A(t), must be chosen sufficiently larger to
reduce carrier leakage. However, as A(t) increases, the input
port of mixers becomes more nonlinear. A compromise is
therefore necessary.
( )
2 2
1 2
2
Relative Carrier Leakage OS OSV V
A t
+
=
• Use BB offset control to reduce
the carrier leakage. During
carrier leakage cancellation, the
BB processor produces a zero
output so that the detector
measures only the leakage. Thus,
the loop can use the DACs to
drive the leakage toward zero.

Transmitter Linearity (I)
• Upconversion mixers in TX sense no interferers, however,
excessive nonlinearity in the mixer BB port can corrupt the
signal or raise the adjacent channel power.
• In most cases, as the BB signal swings increase, the PA output
begins to compress before the mixer nonlinearity manifests
itself.
• Power back-off is required for
variable envelope signal to avoid
spectrum regrowth at PA output.
1-dB Compression Point
outV
inV
0V
t

Transmitter Linearity (II)
• In the TX chain, the signal may experience compression in any
of the stages. Since the largest voltage swing occurs at the
output of the PA, this stage dominates the compression of the
TX. In a good design, the preceding stages must remain well
below compression as the PA output approaches P1dB. To
ensure this, we must maximize the PA gain and minimize the
output swing of the predriver and the stage preceding it.
cos ctω
sin ctω
,BB IV
,BB QV
PAPredriver
XV drV outV
XV
drV
outV
BBV

Oscillator Pulling
• The PA output (very large swing) would couple to various
parts of the system through the silicon substrate, package
parasitics, and traces on the printed-circuit board. Thus, it is
likely that an appreciable fraction of PA output couples to the
LO to pull the oscillator.
outφLO
ω∆
LOω
ω
LOω
ω
ω∆
PA
LO
I
LOω
Q
Output
Spectrum
injω ω

Avoid LO Pulling (I)
• Most of today’s DCTs avoid an oscillator frequency to the PA
output frequency by using frequency division and mixing.
• Since the PA nonlinearity produces a finite amount of power at
the second harmonics of the carrier, the LO may still be pulled
by using the following architecture.
• Very high speed divider is needed, but even a substantial effort
on divider design to enable this architecture is well justified.
I
2LO cω ω=
Q
LO 2÷
PA
cω 2 cω
ω

Avoid LO Pulling (II)
• Use a frequency double is possible to avoid LO pulling, but
the doubler typically dose not provide quadrature phases,
necessitating additional quadrature generation stages such as
the poly phase filter.
• Advantage: no harmonic can pull the LO.
• Disadvantage: the doubler and polyphase filter suffer from a
high loss, requiring the use of power-hungry buffers.
I
2
c
LO
ω
ω =
Q
LO 2X Polyphase
Filter
PA
cω
ω

Avoid LO Pulling (III)
• Use of mixing to avoid LO pulling.
• It is difficult to remove the unwanted carrier by means of
filtering because the two frequencies are only differ by only a
factor of 3. Even the filter is applied, the unwanted sideband
would corrupt other channels or bands.
1
2
ω
LO
2÷
1ω
1
2
ω 13
2
ω
ω
PA
Quadrature
Upconverter
( ),BB Ix t
( ),BB Qx t
chI chQ
LO
1
2
ω 13
2
ω
ω
1
2
ω 13
2
ω
ω

Suppress Unwanted Sideband (I)
• Use the single-sideband (SSB) mixing technique to suppress
the unwanted sideband instead of filtering.
• The harmonics of the input frequencies also corrupt the output
of an SSB mixer.
2cos tω
2sin tω
1cosA tω
1sinA tω
outV
1ω
2ω
outV
( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = +
Symbol of a SSB mixer
2ω 1ω1 2ω ω−0 2 13ω ω− 1 2ω ω+ 1 23ω ω−
ω

Suppress Unwanted Sideband (II)
• For use in a DCT, the SSB mixer must provide the quadrature
phases of the carrier. This is accomplished by nothing that
2cos tω
2sin tω
1cos tω
1sin tω
2cos tω
2sin tω
+
−
+
+
( )1 2sin tω ω+
( )1 2cos tω ω+
( )1 2 1 2 1 2sin cos cos sin sint t t t tω ω ω ω ω ω− = +
( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = +
SSB mixer providing quadrature outputs

Suppress Unwanted Sideband (III)
• Carrier provided by SSB mixing for a DCT.
• While suppressing the carrier sideband at ω1/2, this
architecture presents two drawbacks: (1) the spurs at 5ω1/2 and
other harmonic-related frequencies prove troublesome, and (2)
the LO must provide quadrature phases, a difficult issue.
( ),BB Ix t
( ),BB Qx t
12
3
ω 1
3
ωI/QI/Q
LO 2÷
PA
1ω
DCT using SSB mixing in LO path

Heterodyne Transmitter
• Another approach to avoiding injection pulling involves
performing the signal upconversion in two steps so that the
LO frequency remains far from the PA spectrum.
• Smaller I/Q mismatch
1sin tω
1cos tω
2cos tω
I
BPF
PA
Q
1ω
ω
2ω
ω
1 2ω ω+
ω

Sliding-IF TX
• The carrier frequency is equal to 3ω1/2.
1ω
PA
BPF
LO2÷
I
Q
1
1
2
ω 1
3
2
ω
RF Mixer

Carrier Leakage
• The dc offsets in BB yield a component at ω1/2 at the output of
the quadrature upconverter, and the dc offset at the input of the
RF mixer produces another component at ω1. The former can
be minimized, and the latter (lower sideband) at ω1/2 must be
removed by filtering. The leakage at ω1 is closer to the upper
sideband than the lower sideband is, but it is also much smaller
than the lower sideband. Thus, the filter following the RF
mixer must be designed to attenuate both to acceptably low
levels.
1
2
ω
IF Output
1
2
ω 13
2
ω
1ω
ω
RF Output
Carrier leakage in heterodyne TX

Mixing Spurs (I)
• Heterodyne TX displays various mixing spurs that must be
managed properly. The spurs arise from the mechanism with
1st LO and 2nd LO.
1
2
ω
+
IF Output
13
2
ω
+ 15
2
ω
+015
2
ω
− 13
2
ω
− 1
2
ω
−
ω
1
2
ω
+ 13
2
ω
+ 15
2
ω
+0 17
2
ω
+13
2
ω
− 1
2
ω
−
ω
RF Output
1
2
ω
+ 13
2
ω
+17
2
ω
− 015
2
ω
− 13
2
ω
− 1
2
ω
−
ω
LO mixed with 2LO, 5LO
IF mixed with 2nd LO

Mixing Spurs (II)
• Effect of harmonics of 2nd LO on TX output. Upon mixing
with +3ω1, the IF sideband at −3ω1/2 is translated to +3ω1/2,
thereby corrupting the wanted sideband (if the modulation is
asymmetric). Similarly, the IF sideband at −5ω1/2 is mixed
with +5ω1 and falls atop the desired signal.
13
2
ω
+
ω
13
2
ω
−
015
2
ω
−
13
2
ω
+
ω
0

Reduce Unwanted Components
• Use of BB quadrature SSB mixing and IF SSB mixing to
reduce the unwanted component.
( ) ( ), cosBB Ix t A t θ=
( ) ( ), sinBB Qx t A t θ=
+
−
+
+
1ω
LO2÷
PA
RF SSB Mixer
I
Q
13
2
ω

OOK Transceivers
• On-off keying (OOK) modulation is a special case of ASK
where the carrier amplitude is switched between zero and
maximum.
• Less bandwidth-efficient as unshaped binary pulses modulated
on one phase of the carrier occupy a wide spectrum.
LO
PA
LO
PA
LNA
Envelope
Detector
Direct LO switching PA switching
OOK RX
OOK TX

Open-loop Modulation
• Open-loop modulation based-on a frequency synthesizer (or
phase-locked loop).
• Wideband (high data rate).
• Poor accuracy due to VCO frequency drifting.
reff
DAC
VCO
PFD Loop Filter
Div-by-N
[ ]BBs n
( )BBs t
( )ms t

Closed-loop Modulation (I)
• Closed-loop modulation based-on a frequency synthesizer (or
phase-locked loop).
• Narrowband (low data rate).
• Good frequency accuracy.
• No DACs required.
∆ − ∑
VCO
PFD Loop Filter
[ ]BBs n
/ 1N N÷ +
( )ms t
reff
Modulator

Closed-loop Modulation (II)
• Use the compensated filtering to increase the data rate.
∆ − ∑
VCO
PFD Loop Filter
[ ]BBs n
/ 1N N÷ +
( )ms t
reff
Modulator
Compensated
Filter

Closed-loop Modulation (III)
• Use the two-point ∆-Σ modulation to increase the data rate.
∆ − ∑
Two-point
VCO
PFD Loop Filter
[ ]BBs n / 1N N÷ +
( )ms t
reff
Modulator
DAC

Envelope
Detector
Envelope Following/Tracking Transmitter
• Dynamically adjusting bias to improve efficiency.
( )BBA t′
( )ms t
Linear PA
Antenna
Matching
( )BBA t
I/Q Modulator
Amplitude
Modulator/
Regulator
( )I t
cos ctω
sin ctω−
( )Q t
106/110

Polar Transmitter (I)
• Envelope Elimination and Restoration Scheme (Kahn EER TX,
1952):
Envelope
Detector ( )BBA t′
( )ms t
Switching-mode
PA
Antenna
Matching
( )BBA t
I/Q Modulator
Amplitude
Modulator/
Regulator
( )I t
cos ctω
sin ctω−
( )Q t
Limiter
107/110

Polar Transmitter (II)
• Polar Transmitter
( )BBA t
cos ctω
( )ms t
Switching-mode
PA
Antenna
Phase
Modulator
Matching
( )BBA t
( )BB tφ
Baseband
Processor
Amplitude
Modulator
( )
{ }2
Re c BBj f t t
e
π φ+  
• Linear modulator to generate PM signal
• Frequency synthesizer or PLL-based PM modulator
108/110

Polar Transmitter (III)
• Hybrid Quadrature and Polar Modulation TX (HQPM-TX):
Baseband
Processor
( )BBA t′
( )ms t
Switching-mode
PA
Antenna
Matching
( ),BB DSMA t
I/Q Modulator
Amplitude
Modulator/
Class-S
( )I t
cos ctω
sin ctω−
( )Q t
109/110

Summary
• In this chapter, many receiver and transmitter architectures
were introduced. For receiving or transmitting, there are two
main categories including heterodyne and direct conversion
architectures.
• For these transceivers, the modulation and demodulation can
be classified as “I/Q” and “polar” schemes. I/Q modulator is
an universal modulator with high linearity and signal quality,
and the polar modulator is adopted for improving power
efficiency. I/Q demodulator is the conventional scheme to
demodulate signals, and the polar demodulator is proposed for
low-cost and low-power applications.

RF Module Design - [Chapter 4] Transceiver Architecture

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RF Module Design - [Chapter 4] Transceiver Architecture